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Kingbor Technology Co.,Ltd
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KB3302
2Amp, 2MHz Step-up Switching regulator with Soft-Start
FEATURES

DESCRIPTION
The KB3302 is a high-frequency current-mode step-up switching regulator with an integrated 2A power transistor. Its high switching frequency (programmable up to 2MHz) allows the use of tiny surface-mount external passive components. Programmable soft-start eliminates high inrush current during start-up. The internal switch is rated at 32V making the converter suitable for high voltage applications such as Boost, SEPIC and Flyback. The operating frequency of the KB3302 can be set with an external resistor. The ability to set the operating frequency gives the KB3302 design flexibilities. A dedicated COMP pin allows optimization of the loop response. The KB3302 is available in thermally enhanced 8-Pin MSOP packages.
Up to 95% Efficiency TDB uA No Load Current 1000mA Output Current 1.5V to 16V Input Voltage Range Programmable switching frequency up to 2MHz Output voltage up to 32V Constant switching frequency current-mode control 1.23V Reference Allows Low Output Voltages Shutdown Mode Draws 10 A Supply Current Low saturation voltage switch: 220mV at 2A Overtemperature Protected,Soft-Start function 8-Pin MSOP Packages
APPLICATIONS

Flat screen LCD bias supplies TFT bias supplies XDSL power supplies Medical equipment Digital video cameras Portables devices White LED power supplies
TYPICAL APPLICATION
VIN = 3.3V TO 4.2V C1 4.7F L1 3.3H
1N5819
KB3302 Efficiency
VOUT 5.0V 1000mA
95 VOUT = 5V 90 1.2MHz VIN = 4.2V
6
VIN
5
SW R1 300k
85
3 7
KB3302
SS GND
MSOP8
COMP ROSC
1
R3 17.4k 1nF R2 100k
Efficiency (%)
SHDN
FB
2
C3 22 F
80 75 70 65 60 VIN = 3.6V
100nF
4
8
R3 10.7k
VIN = 2.6V
L1: Sumida CR43
55 50 0.001
Figure 1. 1.2MHzAll Ceramic Capacitor Single L i-ion Cell to 5V Boost Converter.
0.010
0.100
1.000
Load Current (A)
1
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KB3302
Peak SW Sink and Source Current ........................ 2A Operating Temperature Range (Note 2) .. - 40C to 85C Junction Temperature (Note 3) ............................ 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C ORDER PART NUMBER
8 SS 7 ROSC
ABSOLUTE MAXIMUM RATINGS (Note 1)
Input Supply Voltage .................................. - 0.3V to 18V SHDN, V FB Voltages .................................. - 0.3V to 5V SW Voltage ................................................ - 0.3V to 32V
PACkAGE/ORDER INFORMATION
TOP VIEW COMP 1 FB 2 SHDN 3 GND GND 4 5 GND 10 SS 9 ROSC 8 VIN 7 SW 6 SW
ORDER PART NUMBER
COMP 1
TOP VIEW
KB3302DD
3000 Units on Tape and Reel
FB 2
KB3302EMS
2500 Units on Tape and Reel
GND
SHDN 3 GND 4 6 VIN 5 SW
DD PACKAGE 10-LEAD (3mm x 3mm) PLASTIC DFN EXPOSED PAD IS PGND (PIN 11) MUST BE CONNECTED TO GND
DD PART MARKING
8-LEAD PLASTIC MSOP EXPOSED PAD IS PGND MUST BE CONNECTED TO GND TJMAX = 125C, JA = 45C/W, JC = 10C/W
EMS PART MARKING
TJMAX = 125C, JA = 45C/W, JC = 10C/W
ELECTRICAL CHARACTERISTICS
Unless specified: VIN = 2V, SHDN = 1.5V, ROSC = 7.68k, -40C < T A = TJ < 85C
Parameter Undervoltage Lockout Threshold Maximum Operating Voltage Feedback Voltage Feedback Voltage Line Regulation FB Pin Bias Current Error Amplifier Transconductance Error Amplifier Open-Loop Gain COMP Source Current COMP Sink Current VIN Quiescent Supply Current VIN Supply Current in Shutdown Switching Frequency Maximum Duty Cycle Minimum Duty Cycle Switch Current Limit Switch Saturation Voltage
Test Conditions
Min
Typ 1.3
Max 1.4 16
Unit V V V V %
TA = 25C -40C < TA < 85C 1.5V < VIN < 16V
1.224 1.217
1.242
1.260 1.267
0.01 40 60 49 80
nA -1 dB A A
VFB = 1.1V VFB = 1.4V VSHDN = 1.5V, VCOMP = 0 ( Not Switching ) VSHDN = 0 1.3 85
5 5 1.1 10 1.5 90 0 2 2.8 220 350 1.6 18 1.7
mA A MHz % % A mV
ISW = 2A
2
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KB3302
ELECTRICAL CHARACTERISTICS
Unless specified: VIN = 2V, SHDN = 1.5V, ROSC = 7.68k, -40C < T A = TJ < 85C
Parameter Switch Leakage Current Shutdown Threshold Voltage Shutdown Pin Current Soft-Start Charging Current Thermal Shutdown Temperature Thermal Shutdown Hysteresis
Test Conditions VSW = 5V
Min
Typ 0.01
Max 1 1.18
Unit A V A
1.02 VSHDN = 1.2V VSHDN = 0 VSS = 0.3V
1.1 -4.6 0 1.5 160 10
0.1
A A C C
TYPICAL PERFORMANCE CHARACTERISTICS
VIN Current vs SHDN Pin Voltage
1.2 VIN = 2V 1
0.08 0.1 VIN = 2V
VIN Current vs SHDN Pin Voltage
-3
Shutdown Pin Current vs Temperature
V SHDN = 1.25V
VIN Current (mA)
VIN Current (mA)
0.06
Current (A)
0.8 0.6 0.4 0.2 0 0 0.5 1
125C
25C
-4
VIN = 2V
0.04
-5
VIN = 12V
-40C
125C -40C
0.02
0
-6
1.5
0
0.2
0.4
0.6
0.8
1
1.2
-50
-25
0
25
50
75
100
125
SHDN Voltage (V)
SHDN Voltage (V)
Temperature (C)
Soft-Start Charging Current vs Temperature
2
Transconductance vs Temperature
80
V SS = 0.3V
VIN = 2V Transconductance ( )
70
-1
1.8
Current (A)
1.6
60
1.4
50
1.2
40
1 -50 -25 0 25 50 75 100 125
30 -50 -25 0 25 50 75 100 125
Temperature (C)
Temperature (C)
3
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Switching Frequency vs Temperature
1.7
TYPICAL PERFORMANCE CHARACTERISTICS
Feedback Voltage vs Temperature
1.3
100
ROSC vs Switching Frequency
ROSC = 7.68K
Feedback Voltage (V)
1.6
1.25
VIN = 2V ROSC (K ) 25C
10
Frequency (MHz)
VIN = 12V
1.5
VIN = 2V
1.2
1.4
1.15 -50 -25 0 25 50 75 100 125
1 0.0 0.5 1.0 1.5 2.0 2.5 3.0
1.3 -50 -25 0 25 50 75 100 125
Temperature (C)
Frequency (MHz)
Temperature (C)
Switch Saturation Voltage vs Switch Current
400
3
Switch Current Limit vs Temperature
1.5
Minimum VIN vs Temperature
VCESAT (mV)
85C
200
2.6
Input Voltage (V)
-50 -25 0 25 50 75 100
Current Limit (A)
300
25C
2.8
1.4
1.3
-40C
100
2.4
1.2
2.2
1.1
0 0 0.5 1 1.5 2 2.5 3
2
1 -50 -25 0 25 50 75 100 125
Switch Current (A)
Temperature (C)
Temperature (C)
VIN Quiescent Current vs Temperature
1.3
VIN Current in Shutdown vs Input Voltage
50
1.20
Shutdown Threshold vs Temperature
VIN = 2V
Not Switching
1.2
40
Shutdown Threshold (V)
VSHDN = 0
VIN Current (mA)
1.1
VIN Current ( A)
VIN = 16V
-40C
30
1.15
125C
20
1.10
1
VIN = 2V
0.9
1.05
10
0.8 -50 -25 0 25 50 75 100 125
0 0 5 10 15 20
1.00 -50 -25 0 25 50 75 100 125
Temperature (C)
Input Voltage (V)
Temperature (C)
4
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KB3302
PIN FUNCTIONS
Pin 1 2 Pin Name COMP FB Pin Function The output of the internal transconductance error amplifier. This pin is used for loop compensation. The inverting input of the error amplifier. Tie to an external resistive divider to set the output voltage. Shutdown Pin. The accurate 1.1V shutdown threshold and the 4.6uA shutdown pin current hysteresis allow the user to set the undervoltage lockout threshold and hysteresis for the switching regulator. Pulling this pin below 0.1V causes the converter to shut down to low quiescent current. Tie this pin to IN if the UVLO and the shutdown features are not used. This pin should not be left floating. Ground. Tie to the ground plane. Collector of the internal power transistor. Connect to the boost inductor and the rectifying diode. Power Supply Pin. Bypassed with capacitors close to the pin. A resistor from this pin to the ground sets the switching frequency. Soft-Start Pin. A capacitor from this pin to the ground lengthens the start-up time and reduces startup current. The exposed pad must be soldered to the ground plane on the PCB for good thermal conduction.
3
SHDN
4 5 6 7 8
GND SW IN ROSC SS Exposed Pad
SIMPLIFIED BLOC DIAGRAM
IN 6 4.6A SW 5
SHDN 3
+
CMP
1.1V VOLTAGE REFERENCE THERMAL SHUTDOW N ENABLE CLK
INTERNAL SUPPLY
REG
1.242V FB 2 COMP 1 SS 8
+
-
EA REG 1.5A
PWM
R Q S
+
+
ILIM I-LIMIT
R SENSE
REG_GOOD ENABLE
+ +
+
ISEN
4 GND
ROSC 7
CLK
OSCILLATOR
SLOP E COMP
5
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KB3302
APPLICATIONS INFORMATION
Setting the Output Voltage An external resistive divider R1 and R2 with its center tap tied to the FB pin (Figure 4) sets the output voltage.
V R1 = R2 OUT - 1 1.242V
VOUT
OPERATION
The KB3302 is a programmable constant-frequency peak current-mode step-up switching regulator with an integrated 2A power transistor. Referring to the block diagrams in Figures 2 and 3, the power transistor is switched on at the trailing edge of the clock. Switch current is sensed with an integrated sense resistor. The sensed current is summed with the slope-compensating ramp before compared to the output of the error amplifier EA. The PWM comparator trip point determines the switch turn-on pulse width. The current-limit comparator ILIM turns off the power switch when the switch current exceeds the 2.8A current-limit threshold. ILIM therefore provides cycle-by-cycle current limit. Current-limit is not affected by slope compensation because the current comparator ILIM is not in the PWM signal path. Current-mode switching regulators utilize a dual-loop feedback control system. In the KB3302 the amplifier output COMP controls the peak inductor current. This is the inner current loop. The double reactive poles of the output LC filter are reduced to a single real pole by the inner current loop, easing loop compensation. Fast transient response can be obtained with a simple Type-2 compensation network. In the outer loop, the error amplifier regulates the output voltage.
(1)
R1 40nA 2
KB3302
FB
R2
Figure 4. The Output Voltage is set with a Resistive Divider
The input bias current of the error amplifier will introduce an error of:
VOUT 40nA (R1 // R2 )100 = % VOUT 1.242V
(2)
The percentage error of a VOUT = 5V converter with R1 = The switching frequency of the KB3302 can be programmed 100K and R2 = 301K is up to 2MHz with an external resistor from the ROSC pin to the ground. For converters requiring extreme duty VOUT 40nA (100K // 301K )100 = = 0.24% cycles, the operating frequency can be lowered to 1.242V VOUT maintain the necessary minimum on time or the minimum off time. Operating Frequency and Efficiency The KB3302 requires a minimum input of 1.4V to operate. A voltage higher than 1.1V at the shutdown pin enables Switching frequency of KB3302 is set with an external the internal linear regulator REG in the KB3302. After VREG resistor from the ROSC pin to the ground. A graph showing becomes valid, the soft-start capacitor is charged with a the relationship between ROSC and switching frequency is 1.5A current source. A PNP transistor clamps the output given in the "Typical Characteristics". of the error amplifier as the soft-start capacitor voltage rises. Since the COMP voltage controls the peak inductor High frequency operation reduces the size of passive current, the inductor current is ramped gradually during components but switching losses are higher. The efficiencies soft-start, preventing high input start-up current. Under of 5V to 12V converters operating at 700KHz, 1.35MHz fault conditions (VIN<1.4V or over temperature) or when and 2MHz are shown in Figure 1(b). The peak efficiency the shutdown pin is pulled below 1.1V, the soft-start of the KB3302 appears to decrease 0.5% for every capacitor is discharged to ground. Pulling the shutdown 100KHz increase in frequency. pin below 0.1V reduces the total supply current to 10A.
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KB3302
It is worth noting that IOUTMAX is directly proportional to the
APPLICATIONS INFORMATION
Duty Cycle The duty cycle D of a boost converter is:
VIN 1- VOUT + VD D= V 1 - CESAT VOUT + VD
VIN ratio V . Equation (4) over-estimates the maximum OUT
output current at high frequencies (>1MHz) since switching losses are neglected in its derivation. Nevertheless it is a useful first-order approximation. (3) Using VCESAT = 0.3V, VD = 0.5V and ILIM = 2A in (3) and (4), the maximum output currents for three VIN and VOUT combinations are shown in Table 1.
where VCESAT is the switch saturation voltage and VD is voltage drop across the rectifying diode. Maximum Output Current
VIN ( V )
VOUT ( V ) 12 5 12
D 0.820 0.423 0.615
IOUTMAX ( A ) 0.35 1.14 0.76
In a boost switching regulator the inductor is connected to the input. The DC inductor current is the input current. When the power switch is turned on, the inductor current flows into the switch. When the power switch is off, the inductor current flows through the rectifying diode to the output. The output current is the average diode current. The diode current waveform is trapezoidal with pulse width (1 - D)T (Figure 5). The output current available from a boost converter therefore depends on the converter operating duty cycle. The power switch current in the KB3302 is internally limited to 2A. This is also the maximum inductor or the input current. By estimating the conduction losses in both the switch and the diode, an expression of the maximum available output current of a boost converter can be derived:
ILIM VIN D VD - D(VD - VCESAT ) 1 - 45 - VOUT VIN
2.5 3.3 5
Table 1. Calculated Maximum Output Current [ Equation (4)]
Considerations for High Frequency Operation The operating duty cycle of a boost converter decreases as VIN approaches VOUT. The PWM modulating ramp in a current-mode switching regulator is the sensed current ramp of the control switch. This current ramp is absent unless the switch is turned on. The intersection of this ramp with the output of the voltage feedback error amplifier determines the switch pulse width. The propagation delay time required to immediately turn off the switch after it is turned on is the minimum switch on time. Regulator closed-loop measurement shows that the KB3302 has a minimum on time of about 150ns at room temperature. The power switch in the KB3302 is either not turned on at all or for at least 150ns. If the required switch on time is shorter than the minimum on time, the regulator will either skip cycles or it will start to jitter. Example: Determine the maximum operating frequency of a Li-ion cell to 5V converter using the KB3302. Assuming that VD=0.5V, VCESAT=0.3V and VIN=2.6 - 4.2V, the minimum duty ratio can be found using (3).
4.2 5 + 0.5 = 0.25 = 0.3 1- 5 + 0 .5 1-
IOUTMAX =
(4)
where ILIM is the switch current limit.
IIN Inductor Current ON OFF ON Switch Current
Diode Current
DT ON OFF
(1-D)T IOUT ON OFF ON
DMIN
Figure 5. Current Waveforms in a Boost Regulator
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KB3302
D(VIN - VCESAT ) (5) fL where f is the switching frequency and L is the inductance. IL =
Substituting (3) into (5) and neglecting VCESAT ,
APPLICATIONS INFORMATION
The absolute maximum operating frequency of the
DMIN 0.25 = = 1.67MHz . The 150ns 150ns actual operating frequency needs to be lower to allow for modulating headroom.
converter is therefore
V VIN IL = IN 1 - (6) The power transistor in the KB3302 is turned off every fL VOUT + VD switching period for an interval determined by the discharge time of the oscillator ramp and the propagation In current-mode control, the slope of the modulating delay of the power switch. This minimum off time limits (sensed switch current) ramp should be steep enough to the maximum duty cycle of the regulator at a given lessen jittery tendency but not so steep that large flux swing decreases efficiency. Inductor ripple current IL between VOUT 25-40% of the peak inductor current limit is a good switching frequency. A boost converter with high V ratio In compromise. Inductors so chosen are optimized in size requires long switch on time and high duty cycle. If the and DCR. Setting IL = 0.3*(2) = 0.6A, VD=0.5V in (6), required duty cycle is higher than the attainable maximum, V VIN V VIN then the converter will operate in dropout. (Dropout is a = IN 1 - L = IN 1 - (7) 0. 6 f fIL VOUT + VD VOUT + 0.5 condition in which the regulator cannot attain its set output voltage below current limit.) where L is in H and f is in MHz.
The minimum off times of closed-loop boost converters set to various output voltages were measured by lowering their Equation (6) shows that for a given VOUT, IL is the highest input voltages until dropout occurs. It was found that the (VOUT + VD ) . If VIN varies over a wide range, then minimum off time of the KB3302 ranged from 80 to 110ns when VIN = 2 at room temperature. choose L based on the nominal input voltage. Beware of dropout when operating at very low input voltages (1.5-2V) and with off times approaching 110ns. Shorten the PCB trace between the power source and the device input pin, as line drop may be a significant percentage of the input voltage. A regulator in dropout may appear as if it is in current limit. The cycle-by-cycle current limit of the KB3302 is duty-cycle and input voltage invariant and is typically 2.8A. If the switch current limit is not at least 2A, then the converter is likely in dropout. The switching frequency should then be lowered to improve controllability. Both the minimum on time and the minimum off time reduce control range of the PWM regulator. Bench measurement showed that reduced modulating range started to be a problem at frequencies over 2MHz. Although the oscillator is capable of running well above 2MHz, controllability limits the maximum operating frequency. Inductor Selection The inductor ripple current I L of a boost converter operating in continuous-conduction mode is The saturation current of the inductor should be 20-30% higher than the peak current limit (2.8A). Low-cost powder iron cores are not suitable for high-frequency switching power supplies due to their high core losses. Inductors with ferrite cores should be used. Input Capacitor The input current in a boost converter is the inductor current, which is continuous with low RMS current ripples. A 2.2-4.7F ceramic input capacitor is adequate for most applications. Output Capacitor Both ceramic and low ESR tantalum capacitors can be used as output filtering capacitors. Multi-layer ceramic capacitors, due to their extremely low ESR (<5m), are the best choice. Use ceramic capacitors with stable temperature and voltage characteristics. One may be tempted to use Z5U and Y5V ceramic capacitors for output filtering because of their high capacitance and
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KB3302
forward voltages). This is because the diode conduction interval is much longer than that of the transistor. Converter efficiency will be improved if the voltage drop across the diode is lower. The rectifying diodes should be placed close to the SW pins of the KB3302 to minimize ringing due to trace inductance. Surface-mount equivalents of 1N5817, 1N5819, MBRM120 (ON Semi) and 10BQ015 (IRF) are all suitable. Soft-Start Soft-start prevents a DC-DC converter from drawing excessive current (equal to the switch current limit) from the power source during start up. If the soft-start time is made sufficiently long, then the output will enter regulation without overshoot. An external capacitor from the SS pin to the ground and an internal 1.5A charging current source set the soft-start time. The soft-start voltage ramp at the SS pin clamps the error amplifier output. During regulator start-up, COMP voltage follows the SS voltage. The converter starts to switch when its COMP voltage exceeds 0.7V. The peak inductor current is gradually increased until the converter output comes into regulation. If the shutdown pin is forced below 1.1V or if fault is detected, then the soft-start capacitor will be discharged to ground immediately. The SS pin can be left open if soft-start is not required. Shutdown The input voltage and shutdown pin voltage must be greater than 1.4V and 1.1V respectively to enable the KB3302. Forcing the shutdown pin below 1.1V stops switching. Pulling this pin below 0.1V completely shuts off the KB3302. The total VIN current decreases to 10A at 2V. Figure 6 shows several ways of interfacing the control logic to the shutdown pin. Beware that the shutdown pin is a high impedance pin. It should always be driven from a lowimpedance source or tied to a resistive divider. Floating the shutdown pin will result in undefined voltage. In Figure 6(c) the shutdown pin is driven from a logic gate whose VOH is higher than the supply voltage of the KB3302. The diode clamps the maximum shutdown pin voltage to one diode voltage above the input power supply.
APPLICATIONS INFORMATION
small sizes. However these types of capacitors have high temperature and high voltage coefficients. For example, the capacitance of a Z5U capacitor can drop below 60% of its room temperature value at -25C and 90C. X5R ceramic capacitors, which have stable temperature and voltage coefficients, are the preferred type. The diode current waveform in Figure 5 is discontinuous with high ripple-content. In a buck converter the inductor ripple current IL determines the output ripple voltage. The output ripple voltage of a boost regulator is however much higher and is determined by the absolute inductor current. Decreasing the inductor ripple current does not appreciably reduce the output ripple voltage. The current flowing in the output filter capacitor is the difference between the diode current and the output current. This capacitor current has a RMS value of:
IOUT
VOUT -1 VIN
(8)
If a tantalum capacitor is used, then its ripple current rating in addition to its ESR will need to be considered. When the switch is turned on, the output capacitor supplies the load current IOUT (Figure 5). The output ripple voltage due to charging and discharging of the output capacitor is therefore:
VOUT =
IOUTDT COUT
(9)
For most applications, a 10-22F ceramic capacitor is sufficient for output filtering. It is worth noting that the output ripple voltage due to discharging of a 10F ceramic capacitor (9) is higher than that due to its ESR. Rectifying Diode For high efficiency, Schottky barrier diodes should be used as rectifying diodes for the KB3302. These diodes should have a RMS current rating of at least 1A and a reverse blocking voltage of at least a few Volts higher than the output voltage. For switching regulators operating at low duty cycles (i.e. low output voltage to input voltage conversion ratios), it is beneficial to use rectifying diodes with somewhat higher RMS current ratings (thus lower
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APPLICATIONS INFORMATION
IN KB3302 IN KB3302
SHDN
SHDN
(a)
(b)
VIN 1N4148
IN KB3302
IN KB3302
SHDN
SHDN
(c)
(d)
Figure 6. Methods of Driving the Shutdown Pin (a) Directly Driven from a Logic Gate (b) Driven from an Open-drain N-channel MOSFET or an Open-Collector NPN Transistor (VOL < 0.1V) (c) Driven from a Logic Gate with VOH > VIN (d) Combining Shutdown with Programmed UVLO (See Section Below).
Programming Undervoltage Lockout
VH and VL are therefore: The KB3302 has an internal VIN undervoltage lockout (UVLO) threshold of 1.4V. The transition from idle to switching is abrupt but there is no hysteresis. If the input voltage ramp rate is slow and the input bypass is limited, then sudden turn on of the power transistor will cause a dip in the line voltage. Switching will stop if VIN falls below the internal UVLO threshold. The resulting output voltage rise may be non-monotonic. The 1.1V disable threshold of the KB3302 can be used in conjunction with a resistive voltage divider to raise the UVLO threshold and to add an UVLO hysteresis. Figure 7 shows the scheme. Both VH and VL (the desired upper and the lower UVLO threshold voltages) are determined by the 1.1V threshold crossings,
R VH = 1 + 3 (1.1 V ) R4 VL = VH - VHYS = VH - IHYSR3
(10)
Re-arranging,
R3 =
R4 =
VHYS IHYS
R3 VH -1 1 .1
(11)
(12)
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The turn off voltage is:
VL = VH - VHYS = 2.75 - 0.69 = 2.06 V > 1.4 V .
APPLICATIONS INFORMATION
IN 6/8
Frequency Compensation
I HYS 4.6A
R3
SWITCH CLOSED WHEN Y = "1" SHDN 3 + Y COMPARATOR
Figure 8 shows the equivalent circuit of a boost converter using the KB3302. The output filter capacitor and the load form an output pole at frequency:
p2 = -
2IOUT 2 =- VOUTC2 ROUTC2
(13)
R4
1.1V
where C2 is the output capacitor and ROUT = equivalent load resistance.
VOUT is the IOUT
KB3302
Figure 7. Programmable Hysteretic UVLO Circuit
The zero formed by C2 and its equivalent series resistance (ESR) is neglected due to low ESR of the ceramic output capacitor. There is also a right half plane (RHP) zero at angular frequency:
Z 2 = ROUT (1 - D )2 L
with VL > 1.4 V . Example: Increase the turn on voltage of a VIN = 3.3V boost converter from 1.4V to 2.75V. Using VH = 2.75V and R4 = 100K in (12),
R3 = 150K .
(14)
z2 decreases with increasing duty cycle D and increasing IOUT. Using the 5V to 12V boost regulator (1.35MHz) in Figure 1(a) as an example,
The resulting UVLO hysteresis is:
VHYS = IHYSR3 = 4.6A * 150K = 0.69V .
ROUT
5V = 6.8 0.74 A
I OUT VOUT ESR R1 C2 R OUT
V IN
POWER STAGE C5
COMP R3 C6 C4 RO
Gm
+
FB
1.242V VOLTAGE REFERENCE
R2
Figure 8. Simplified Block Diagram of a Boost Converter
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p1 = - 1 1 =- RO C 4 4.7M * 820pF
APPLICATIONS INFORMATION
5 12 + 0.5 = 0.62 D= 0 .3 1- 12 + 0.5 1-
= -260 rads -1 = -41Hz
Therefore
p 2 2 = 29.4Krads-1 = 4.68KHz (6.8 ) * (10F )
6.8 * (1 - 0.62)2 = 209 Krads -1 = 33.3KHz 4.7H
C4 and R3 also forms a zero with angular frequency:
Z1 = -
1 1 =- R 3C 4 30.9K * 820pF
= -39.5 Krads -1 = -6.3 KHz
The poles p1, p2 and the RHP zero z2 all increase phase shift in the loop response. For stable operation, the overall loop gain should cross 0dB with -20dB/decade slope. Due to the presence of the RHP zero, the 0dB crossover frequency
and
Z 2
The spacing between p2 and z2 is the closest when the converter is delivering the maximum output current from the lowest VIN. This represents the worst-case compensation condition. Ignoring C5 and C6 for the moment, C4 forms a low frequency pole with the equivalent output resistance RO of the error amplifier:
Amplifier Open Loop Gain 49dB RO = = = 4.7M Transconduc tan ce 60 -1
z2 . Placing z1 near p2 nulls its 3 effect and maximizes loop bandwidth. Thus
should not be higher than
R 3C 4 VOUT C2 2IOUT (MAX )
(15)
R3 determines the mid-band loop gain of the converter. Increasing R3 increases the mid-band gain and the crossover
GND
C3 R3 C4 C6 R2 U1 C1 R4
SHDN
R1
C5 C2 D1
L1
VOUT
VIN
Figure 9. Suggested PCB Layout for the KB3302. Notice that there is no via directly under the device. All vias are 12mil in diameter.
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KB3302
the size of the loop formed by these components should be minimized. Since the power switch is integrated inside the KB3302, grounding the output filter capacitor next to the KB3302 ground pin minimizes size of the high di/dt current loop. The input bypass capacitors should also be placed close to the input pins. Shortening the trace at the SW node reduces the parasitic trace inductance. This not only reduces EMI but also decreases the sizes of the switching voltage spikes and glitches. Figure 9 shows how various external components are placed around the KB3302. The frequency-setting resistor should be placed near the ROSC pin with a short ground trace on the PC board. These precautions reduce switching noise pickup at the ROSC pin. To achieve a junction to ambient thermal resistance (JA) of 40C/W, the exposed pad of the KB3302 should be properly soldered to a large ground plane. Use only 12mil diameter vias in the ground plane if necessary. Avoid using larger vias under the device. Molten solder may seep through large vias during reflow, resulting in poor adhesion, poor thermal conductivity and low reliability.
APPLICATIONS INFORMATION
frequency. However it reduces the phase margin. The values of R3 and C4 can be determined empirically by observing the inductor current and the output voltage during load transient. Compensation is optimized when the largest R3 and the smallest C4without producing ringing or excessive overshoot in its inductor current and output voltage are found. C5 adds a feedforward zero to the loop response. In some cases it improves the transient speed of the converter. C6 rolls off the gain at high frequency. This helps to stabilize the loop. C5 and C6 are often not needed. Board Layout Considerations In a step-up switching regulator, the output filter capacitor, the main power switch and the rectifying diode carry switched currents with high di/dt. For jitter-free operation,
Typical Application Circuits
VIN 5V 6 OFF ON 3 C1 2.2F 8 IN SHDN KB3302 SS GND C3 47nF 4 COMP ROSC 7 R4 C4 1 R3 C6 R2 20K 5 SW FB 2 C2 10F L1 D1 VOUT 12V, 0.7A R1 174K 10BQ015
Efficiency
95 10.5H, 700KHz 90 85 4.7H, 1.4MHz
Efficiency (%)
80 3.3H, 2MHz 75 70 65 60
All Capacitors are Ceramic. MSOP-8 Pinout
f / MHz 0.7 1.35 2 R3 / K 22.1 30.9 63.4 R4 / K 22.1 9.31 4.75 C4 / pF 2200 820 470 C6 / pF 22 L1 / H 10.5 (Falco D08019) 4.7 (Falco D08017) 3.3 (Coilcraft DO1813P)
55 50 0.0 0.1 0.2 0.3 0.4
VIN = 5V VOUT = 12V 0.5 0.6 0.7
Load Current (A)
Figure 10(a). 1.35 MHz All Ceramic Capacitor 5V to 12V Boost Converter. Pinout Shown is for MSOP-8
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KB3302
PACAGE DESCRIPTION
Efficiency
95
2.6 - 4.2V
L1 1.8H 6 OFF ON 3 IN SHDN KB3302 8 SS GND C3 47nF 4 COMP ROSC 7 R4 10.7K 1 5 SW FB 2
D1
VOUT = 5V
VOUT 5V, 0.8A
Efficiency (%)
90 85 80 75 70 65
1.2MHz VIN = 4.2V
10BQ015 R1 301K
1-CELL LI-ION
C1 2.2F
C2 10F R3 17.4K C4 1nF R2 100K
60 VIN = 3.6V 55 50 0.001 VIN = 2.6V
0.010
0.100
1.000
L1: Sumida CR43 Figure 11(a). 1.2 MHz All Ceramic Capacitor Single Li-ion Cell to 5V Boost Converter.
Load Current (A)
Figure 11(b). Efficiency of the Single Li-ion Cell to 5V Boost Converter in Figure 11(a).
4-CELL 3.6 - 6V
L1 4.9H 6 OFF ON 3 C1 2.2F 8 IN SHDN KB3302 SS GND C3 47nF 4 COMP ROSC 7 R4 7.68K 1 5 SW FB 2
C6
D1
VOUT 5V
2.2F
10BQ015 C5 47pF R1 60.4K
C2 10F R3 20K C4 560pF L2 4.9H R2 20K
L1 and L2: Coiltronics CTX5-1 Figure 12(a). 1.5 MHz All Ceramic Capacitor 4-Cell to 5V SEPIC Converter. Pinout Shown is for MSOP-8.
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KB3302
D3 D4 D5 OUT2 23V (10mA) C8 1F C5 0.1F C6 0.1F C7 0.1F
D2
3.3V
L1 2.2H R5 150K 3 6 IN SHDN KB3302 8 R6 100K C3 47nF SS GND 4 COMP ROSC 7 R4 7.68K 1 R3 40.2K C4 820pF 5 SW FB 2
D1
OUT1 8V (0.55A) R1 274K
10BQ015
C1 2.2F
C9 0.1F R2 49.9K
C2 10F
D7
OUT3 -8V (10mA) C10 1F
L1 : Cooper-Bussmann SD25-2R2 D2 - D7 : BAT54S
D6
Figure 13(a). 1.5MHz Triple-Output TFT Power Supply.
- 3.4V to 3.8V + 0.7A (FLASH) 0.2A (TORCH) R6 0.1 R1 698 D2 LXCL-PWF1 D1
2.6 - 4.2V
L1 2.2H SUMIDA CR43 +
10BQ015 + 1/2 LM358
1-CELL LI-ION C1 2.2F
6 OFF ON 3 IN SHDN KB3302 8 SS GND C3 10nF 4
5 SW FB COMP ROSC 7 R4 8.06K C4 10nF R5 10K 2 1 C5 0.1F R6 17.4K
Q1 MMBT3904T
C2 4.7F
R2 43.2K M1 MMBF2201NT1
TORCH FLASH
Figure 14(a). 1.4MHz LuxeonTM Flash White LED Driver for Camera Phones
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KB3302
DIMENSIONS INCHES MILLIMETERS MIN NOM MAX MIN NOM MAX
.043 .006 .000 .037 .030 .009 .015 .009 .003 .114 .118 .122 .114 .118 .122 .193 BSC .026 BSC .068 .076 .080 .016 .024 .032 (.037) 8 0 8 .004 .005 .010 1.10 0.00 0.15 0.75 0.95 0.22 0.38 0.08 0.23 2.90 3.00 3.10 2.90 3.00 3.10 4.90 BSC 0.65 BSC 1.73 1.93 2.03 0.40 0.60 0.80 (0.95) 8 8 0 0.10 0.13 0.25
PACAGE DESCRIPTION - MSOP8
e/2 A N 2X E/2 E1 PIN 1 INDICATOR ccc C 2X N/2 TIPS 12 e B aaa C SEATING PLANE C D A2 A A1 bxN bbb F EXPOSED PAD 0.25 F DETAIL (L1) C A-B D GAGE PLANE L c E D
DIM
A A1 A2 b c D E1 E e F L L1 N 01 aaa bbb ccc
H
01
A
BOTTOM VIEW SIDE VIEW
NOTES: 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). -HTO BE DETERMINED AT DATUM PLANE 2. DATUMS -A- AND -B-
SEE DETAIL
A
3. DIMENSIONS "E1" AND "D" DO NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. 4. REFERENCE JEDEC STD MO-187, VARIATION AA-T.
Land Pattern - MSOP-8L-EDP
F
DIM
(C) G F Z C F G P X Y Z
DIMENSIONS INCHES MILLIMETERS
(.161) .081 .098 .026 .016 .063 .224 (4.10) 2.08 2.50 0.65 0.40 1.60 5.70
P X
NOTES: 1.
THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR COMPANY'S MANUFACTURING GUIDELINES ARE MET.
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KB3511
PACAGE DESCRIPTION - DFN33
A E B
DIM
A A1 A2 b C D E e L N aaa bbb
DIMENSIONS INCHES MILLIMETERS MIN NOM MAX MIN NOM MAX
.031 .039 .000 .002 (.008) .007 .009 .011 .074 .079 .083 .042 .048 .052 .114 .118 .122 .020 BSC .012 .016 .020 10 .003 .004 0.80 1.00 0.00 0.05 (0.20) 0.18 0.23 0.30 1.87 2.02 2.12 1.06 1.21 1.31 2.90 3.00 3.10 0.50 BSC 0.30 0.40 0.50 10 0.08 0.10
E PIN 1 INDICATOR (LASER MARK)
A aaa C A1 C 1 LxN 2 A2 C SEATING PLANE
D
N e bxN bbb CAB
NOTES:
1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS TERMINALS.
Land Pattern - DFN33-10
K
DIM
H C G H K P X Y Z
DIMENSIONS INCHES MILLIMETERS
(.112) .075 .055 .087 .020 .012 .037 .150 (2.85) 1.90 1.40 2.20 0.50 0.30 0.95 3.80
(C)
G
Z
Y X P
NOTES: 1. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR COMPANY'S MANUFACTURING GUIDELINES ARE MET.
Kingbor Technology
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